Methods and apparatus to improve performance and enable fast decoding of transmissions with multiple code blocks

ABSTRACT

A method includes separating resource elements from multiple code blocks into different groups, and decoding the code bits of the resource elements within each group without waiting for a completed reception of a transport block to start decoding. 
     A method includes separating coded bits from multiple code blocks into different groups, and decoding the code blocks containing coded bits within each group. A first CRC is attached to the transport block and a second CRC is attached to at least one code block from the transport block. 
     An improved channel interleaver design method including mapping from coded bits of different code blocks to modulation symbols, and mapping from modulation symbols to time, frequency, and spatial resources, to make sure each code block to get roughly the same level of protection.

CLAIM OF PRIORITY

This application makes reference to, incorporates the same herein, andclaims all benefits accruing under 35 U.S.C. §119 from provisionalapplications earlier filed in the U.S. Patent & Trademark Office on 16Mar. 2007 and there duly assigned Ser. No. 60/918,503, filed on 26 Mar.2007 and assigned Ser. No. 60/920,056, and filed on 27 Mar. 2007 andassigned Ser. No. 60/920,324.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to methods and apparatus for a datatransmission in a communication system, and more specifically, tomethods and apparatus for improving performance of transmission withmultiple code blocks and enabling fast decoding of transmissions withmultiple code blocks in a communication system.

2. Description of the Related Art

Orthogonal Frequency Division Multiplexing (OFDM) is a technology tomultiplex data in frequency domain. Modulation symbols are carried onfrequency sub-carriers and the sub-carriers overlap with each other infrequency domain. The orthogonality is, however, maintained at thesampling frequency in the assumption that the transmitter and receiverhave perfect frequency synchronization. In the case of frequency offsetdue to an imperfect frequency synchronization or due to high mobility,the orthogonality of the sub-carriers at sampling frequencies isdestroyed, resulting in Inter-Carrier-Interference (ICI).

A cylic prefix (CP) portion of the received signal is often corrupted bythe previous Orthogonal Frequency Division Multiplexing (OFDM) symbol ofmultipath fading. When the cylic prefix (CP) portion is sufficientlylong, the received Orthogonal Frequency Division Multiplexing (OFDM)symbol without a cylic prefix (CP) portion should only contain its ownsignal convoluted by the multipath fading channel. The main advantage ofOrthogonal Frequency Division Multiplexing (OFDM) over othertransmission schemes is that Orthogonal Frequency Division Multiplexing(OFDM) demonstrates robustness to compensate for multipath fading.

Single Carrier Frequency Division Multiple Access (SC-FDMA) thatutilizes single carrier modulation and frequency domain equalization, isa technique that has similar performance and complexity to that of anOrthognal Frequency Division Multiplexing Access (OFDMA) system. SingleCarrier Frequency Division Multiple Access (SC-FDMA) is selected as theuplink multiple access scheme in the 3rd Generation Partnership Project(3GPP) Long Term Evolution (LTE). 3GPP LTE is a project within the ThirdGeneration Partnership Project to improve the Universal MobileTelecommunications System mobile phone standard to cope with futurerequirements.

Hybrid Automatic Repeat reQuestion (HARQ) is widely used incommunication systems to combat decoding failure and improvereliability. N-channel synchronous Hybrid Automatic Repeat reQuestion(HARQ) is often used in wireless communication systems because of thesimplicity of N-channel synchronous Hybrid Automatic Repeat reQuestion(HARQ). The synchronous Hybrid Automatic Repeat reQuestion (HARQ) hasbeen accepted as the HARQ scheme for long term evolution (LTE) uplink in3GPP. On the downlink of LTE systems, asynchronous adaptive HARQ hasbeen accepted as the HARQ scheme due to its flexibility and additionalperformance benefits beyond synchronous HARQ.

Multiple antenna communication systems, which are often referred to asMultiple Input Multiple Output (MIMO) systems, are widely used inwireless communication to improve the performance of communicationsystems. In a MIMO system, a transmitter has multiple antennas capableof transmitting independent signals and a receiver is equipped withmultiple receiving antennas. Many MIMO schemes are often used in anadvanced wireless system.

When a channel is favorable, e.g., when the mobile speed is low, it ispossible to use a closed-loop Multiple Input Multiple Output (MIMO)scheme to improve the system performance. In closed-loop MIMO systems,the receivers feed back to the transmitter the channel condition and/orpreferred transmission MIMO processing schemes. The transmitter utilizesthis feedback information, together with other considerations such asscheduling priority, data and resource availability, to jointly optimizethe transmission scheme. A popular closed loop MIMO scheme is calledMIMO preceding. With preceding, the transmit data streams arepre-multiplied by a preceding matrix before being passed on to themultiple transmit antennas.

Another perspective of a Multiple Input Multiple Output (MIMO) system iswhether the multiple data streams for transmission are encodedseparately or encoded together. All the layers for data transmission areencoded together in the Single Codeword (SCW) MIMO system, while all thelayers may be encoded separately in the Multiple Codeword (MCW) MIMOsystem. Both Single User MIMO (SU-MIMO) and Multi-User MIMO (MU-MIMO)are adopted in the downlink of Long Term Evolution (LTE). MU-MIMO isalso adopted in the uplink of Long Term Evolution (LTE), the adoption ofSU-MIMO for Long Term Evolution (LTE) uplink, however, is still underdiscussion.

In a Long Term Evolution (LTE) system, when the transport block islarge, the transport block is segmented into multiple code blocks sothat multiple coded packets can be generated. This break-down oftransport block provides benefits such as enabling parallel processingor pipeline implementation and flexible trade-off between powerconsumption and hardware complexity.

Different modulation schemes, such as Quadrature phase shift keying(QPSK), binary phase shift keying (BPSK), 8 Phase-shift keying (8-PSK),16 Quadrature amplitude modulation (16-QAM), or 64 Quadrature amplitudemodulation (64-QAM) may be used for adaptive modulation and forincreasing the spectral efficiency of modulation. In case of 16-QAMmodulation, quadruples of bits, b0b1b2b3, are mapped to complex-valuedmodulation symbols x=I+jQ. Different modulation positions, however, havedifferent protection levels.

When multiple code blocks are transmitted, the performance of thetransmission is dictated by the code block that has the worstperformance. Channel interleaver, including mapping from coded bits ofdifferent code blocks to modulation symbols, and mapping from modulationsymbols to time, frequency, and spatial resources, needs to be carefullydesigned to make sure that each code block gets roughly the same levelof protection. When multiple code blocks are transmitted, it isbeneficial to allow the receiver to start the decoding of some codeblocks while the receiver is still demodulating modulation symbols forother code blocks. In a long term evolution (LTE) system, this presentsa challenge because the channel estimation performance might bedeleteriously impacted if there are not enough reference signals at thetime of demodulation and decoding.

In order to maintain good channel estimation performance, interpolationof reference signals at selected resource elements located around aresource element to be estimated is often used to obtain channelestimation for the resource element with improved performance. Thishowever, means that the demodulation of the modulation symbol in theresource element to be estimated needs to wait until all the resourceelements selected for estimating the resource element are received. Inother words, if the need for demodulation of the resource element to beestimated occurs before reception of the Orthogonal Frequency DivisionMultiplexing (OFDM) symbol which contains some of or all of the selectedresource elements for estimating the resource element, the channelestimation performance for resource elements may be deleteriouslyaffected.

SUMMARY OF THE INVENTION

It is, therefore, an object of the present invention to provide improvedmethods and apparatus of transmission of signals with multiple codeblocks.

It is another object of the present invention to provide an improveddesign of channel interleavers and improved wireless receivers.

It is another object of the present invention to provide methods andapparatus enabling fast decoding of multiple code blocks whilemaintaining good channel estimation performance.

It is another object of this invention to provide an improved method andan improved apparatus for transmitting data by enabling fast decoding oftransmissions of signals carrying multiple code blocks.

In one embodiment of the present invention, an improved design of achannel interleaver and receiver is provided and a separate codingmethod of multiple code blocks is taken into account in order to improvethe performance. The design for the channel interleaver, including themapping from coded bits of different code blocks to modulation symbols,and the mapping from modulation symbols to time, frequency, and spatialresources, assures that each code block gets roughly the same level ofprotection. On the receiver side, when some code blocks are receivedcorrectly and some are not, the signal of the successfully decoded codeblocks may be reconstructed and cancelled from the received signal.After the cancellation, the receiver may attempt to re-decode the othercode blocks. The interference with other code blocks that are not yetsuccessfully decoded may be therefore greatly reduced, and theprobability that the receiver will be able to decode the other codeblocks may thus be significantly increased.

In one embodiment of the invention prior to transmission, a CRC is addedto each code block to enable error detection for each code block. Afterthe transport block CRC attachment, the bit scrambling, and the codeblock segmentation, a code block CRC is attached to at least one of thecode blocks and the signal is transmitted. Note that if there is onlyone code block in the transport block, the code block CRC may not benecessary. The CRC overhead may be further reduced by only attaching onecode block CRC for multiple code blocks prior to transmission.

In the present invention, a number of steps are provided to be appliedin the improved channel interleaver design.

Step 1

Firstly, for each code block, symbols S, P₁, P₂, contemplate,respectively, the systematic bits, parity bits from encoder 1 of a turboencoder, and parity bits from encoder 2 of the turbo encoder. In oneembodiment of the present invention, the coded bits after the secondrate matching are re-arranged based on code blocks. The re-arranged bitsmay be used to fill up the time-frequency resources, and the modulationpositions in modulation symbols.

Step 2

Secondly, these bits first fill up the space along the dimension offrequency (i.e. sub-carrier) index. Then they fill up the space alongthe dimension of time (i.e. OFDM symbol) index. At last they fill up thespace along the dimension of modulation position index. Other orderingof dimensions is certainly possible and covered by the presentinvention.

Step 3

Thirdly, for each modulation position index and each OrthogonalFrequency Division Multiplexing (OFDM) symbol, the data bits areinterleaved along the frequency dimension. For example, abit-reverse-order (BRO) interleaver or a pruned bit-reverse-orderinterleaver may be used. Or any other interleaver may be used for thispurpose. Sometimes, one or multiple of simplified shuffling patterns maybe used. For example, cyclic shifts, or predeterminedinterleaving/re-arrangement/shuffling/swapping patterns may be used.These patterns may or may not change for each OFDM symbol and/or eachmodulation position index. Sometimes the number of resource elementsavailable in each OFDM symbol may be different due to different amountof puncturing or usage by other channels in these OFDM symbols. In thatcase, interleaver with different sizes may be used on different OFDMsymbols.

Step 4

Fourth, for each modulation position index and each sub-carrier, thedata bits are interleaved along the time dimension. For example, abit-reverse-order (BRO) interleaver or a pruned bit-reverse-orderinterleaver may be used. Or any other interleaver can be used for thispurpose. Sometimes, one or multiple of simplified shuffling patterns canbe used. For example, cyclic shifts, or predeterminedinterleaving/re-arrangement/shuffling/swapping patterns can be used.These patterns may or may not change for each modulation position and/orsub-carrier index. Sometimes, the number of resource elements availableon each sub-carrier index may be different due to different amount ofpuncturing or usage by other channels on this sub-carrier. In that case,interleaver with different sizes may be used on different sub-carriers.

Step 5

Fifth, for each sub-carrier and each OFDM symbol, the data bits areinterleaved along the dimension of modulation position index. Forexample, a bit-reverse-order (BRO) interleaver or a prunedbit-reverse-order interleaver may be used. Or any other interleaver canbe used for this purpose. Sometimes, one or multiple of simplifiedshuffling patterns can be used. For example, cyclic shifts, orpredetermined interleaving/re-arrangement/shuffling/swapping patternscan be used. These patterns may or may not change for each sub-carrierand/or each OFDM symbol. Preferred patterns will be explained later inthe present invention.

Another preferred embodiment of the channel interleaver design consistsof at least one of the above-stated five steps.

The aforementioned embodiments of channel interleaver design may beeasily extended to the case of MIMO transmissions. Suppose multiplelayers are allocated to a MIMO codeword. This scenario may apply to longterm evolution (LTE) systems, e.g., when the SU-MIMO transmission hasrank greater than 1. In this case, a spatial dimension is added in thechannel interleaver design. The space for the coded bits may bedescribed as a four-dimensional space in time, frequency, space andmodulation positions.

In another embodiment of the invention, the aforementioned embodimentsare extended to MIMO transmissions with different spatial dimensions ondifferent resource elements.

In a MIMO system, the rank (number of spatial dimensions, or layers) maybe different on different frequency resource elements. Theaforementioned embodiments may also be extended to transmissions withdifferent modulation order on different resources. For example, if tworesource blocks have very different CQI, the transmitter may usedifferent modulation orders on these two resource blocks. In this case,the design goal of spreading coded bits of each code block as much aspossible over time, frequency, spatial, and modulation positions stillapplies. Special handling needs to be implemented to handle the case ofdifferent spatial dimensions or different modulation orders on differenttime-frequency resources. For example, similar to the resource elementmap, a map can be constructed to include spatial and modulation positiondimension. The layers or the modulation positions that are not availablewill be skipped.

In another embodiment of the invention, systematic bits priority isgiven in mapping coded bits, and modulation symbols formed by thesecoded bits, onto resource elements and spatial dimensions.

The prioritization of systematic bit may also be implemented by definingmultiple regions along the dimension of modulation positions.

In another embodiment of the invention, the coded bits of each codeblock are allocated as uniformly as possible on different modulationpositions. There are various ways to achieve this goal. One approach isto enumerate all the permutation patterns of the modulation positions.

A subset of the permutation patterns may be selected. For example, oneseed permutation pattern with its cyclic shifted versions may be used asone subset of patterns.

Certainly, the selection of a subset of permutation patterns may bevarious and depends on other design objectives. For example, not allcyclic shifts are needed in the selected subset. Cyclic shifts frommultiple seed permutation patterns may be selected.

Different preferred seed permutation patterns, and their cyclic shifts,may be obtained by reading the positions along a circle, starting fromany position and by going either clockwise or counter-clockwise. In thisway, maximum separation of the modulation positions with the same levelof protection is achieved. This method is also applicable to othermodulation orders. Although the seed permutation patterns are generatedin this way, and their cyclic shifts, are preferable, this inventioncertainly covers the application of the modulation positioninterleaving, permutation, shuffling, swapping, re-arranging on resourceelements and/or across retransmissions with any pattern or in anyfashion.

In another embodiment of this invention, an iterative operation isproposed for receiving the multiple code blocks that are multiplexedtogether within modulation symbols. With the aforementioned channelinterleaver design, the coded bits of different code blocks aremultiplexed in the same modulation symbol.

Parallel processing may be also possible in the decoding operation.After the decoding operation, some code blocks may be successfullydecoded while some others are not. In this case, the code blocks ofthose decoded code blocks are reconstructed. Because the coded bits ofthese blocks are multiplexed in the same modulation symbols with thecoded bits of those code blocks that are unsuccessful, the informationof these coded bits are used to help the detection of the coded blocksthat are yet unsuccessful.

In another embodiment of the present invention, a reduced constellationmay improve the detection performance of the transmission.

In another embodiment of the present invention, the iterative operationmay be performed without correctly decoding and re-encoding some of thecode blocks. Instead, reliability of the coded and information bits maybe used to pass through the iterations to improve detection performance.One representation of reliability is called extrinsic information, whichis the new likelihood information about each bit that is passed betweenthe multiple processing blocks within the iterative loop.

In another embodiment of the invention, multiple OFDM symbols in asubframe are separated into numbers of groups with the boundary betweenat least two groups located in the Reference Signal (RS) OFDM symbols,or those OFDM symbols right before or right after the RS OFDM symbols.Each group contains resource elements that will carry coded bits from atleast one code block. The resource elements in each group are contiguousor close to each other in time domain. Therefore, the receiver can startdecoding of at least one code block after receiving all the resourceelements in each group. Different configuration of groups can be used indifferent situations, such as, but not limited to, different UEs,different subframes, different quality of service, etc. withoutdeparting from the spirit of this invention.

In another embodiment of this invention, the groups are defined based oncode blocks instead of resource elements. Each group contains coded bitsof at least one code block and may contain multiple code blocks.

With the group defined in aforementioned embodiments, either based onresource elements or code blocks, the rest of channel interleavingoperations may be defined within each group.

The aforementioned embodiments of channel interleaver design can beextended to the case of MIMO transmissions. When the SU-MIMOtransmission has a transmission rank greater than 1, multiple layers areallocated to a MIMO codeword. In this case, a spatial dimension can beadded to the definition of one group. Therefore, there may be multiplelayers or streams within each group, and there may be multiple groupswithin each MIMO layer or MIMO stream. In a multi-codeword MIMOtransmission, the layers or streams may contain multiple MIMO codewords(CW), each of which carries multiple code blocks and a 24-bit Cyclicredundancy check (CRC). The demodulation of the later groups isparallelized with the decoding of earlier groups. With the help of CRC,the interference from one codeword to another codeword is cancelled bysuccessive interference cancellation.

In another embodiment of this invention, Cyclic redundancy check (CRC)may be added to one or multiple code blocks of a codeword within onegroup. By doing so, the demodulation of the later groups in onecodeword, the decoding of earlier groups in this codeword, thesuccessive interference cancellation, the demodulation of the latergroups in another codeword, and the decoding of earlier groups in theother codeword can all be processed paralleled in one way or another.

In another embodiment of this invention, Cyclic redundancy check (CRC)may be added to the groups of multiple MIMO codewords separately. Inthis embodiment, parallel processing may be enabled even for aniterative receiver.

Several variations and receiver structures may be obtained withoutdeviating from the principle of this invention.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete appreciation of the invention, and many of the attendantadvantages thereof, will be readily apparent as the same becomes betterunderstood by reference to the following detailed description whenconsidered in conjunction with the accompanying drawings in which likereference symbols indicate the same or similar components, wherein:

FIG. 1 is an illustration of an Orthogonal Frequency DivisionMultiplexing (OFDM) transceiver chain having transmitter chain andreceiver chain;

FIG. 2 is a two coordinate illustration of the orthogonality ofOrthogonal Frequency Division Multiplexing (OFDM) theory;

FIG. 3 a is an illustration of an Orthogonal Frequency DivisionMultiplexing (OFDM) symbol in time domain at the transmitter;

FIG. 3 b is an illustration of an Orthogonal Frequency DivisionMultiplexing (OFDM) symbol in time domain at the receiver;

FIG. 4 shows an example of a transceiver chain for Single CarrierFrequency Division Multiple Access (SC-FDMA);

FIG. 5 is an illustration of Hybrid Automatic Repeat reQuestion (HARQ)operation;

FIG. 6 shows an example of a four-channel synchronous Hybrid AutomaticRepeat reQuestion (HARQ);

FIG. 7 is an illustration of a Multiple Input Multiple Output (MIMO)system;

FIG. 8 is an illustration of a Multiple Input Multiple Output (MIMO)preceding process as used in a closed-loop MIMO system;

FIG. 9 is a flow chart of a coding chain for High Speed Data SharedChannel (HS-DSCH) in High Speed Downlink Packet Access (HSDPA);

FIG. 10 is an illustration of the functionality of HS-DSCH HARQ in HighSpeed Downlink Packet Access (HSDPA);

FIG. 11 shows a two dimensional coordinate which shows one illustrationof 16-QAM constellation diagram.

FIG. 12 shows a two dimensional coordinate which shows one illustrationof 64-QAM constellation diagram.

FIG. 13 shows an example of the attachment of code block CRC suitablefor the practice of the principles of one embodiment of the presentinvention;

FIG. 14 illustrates channel interleaver for Orthogonal FrequencyDivision Multiplexing (OFDM) systems suitable for the practice of theprinciples of another embodiment of the present invention;

FIG. 15 illustrates a resource element map for a data transmissionsuitable for the practice of the principles of another embodiment of thepresent invention;

FIG. 16 shows a re-arrangement of coded bits by code blocks after ratematching suitable for the practice of the principles of anotherembodiment of the present invention;

FIG. 17 (a) shows the resources elements represented in one dimensionsuitable for the practice of the principles of another embodiment of thepresent invention;

FIG. 17 (b) shows a time index (OFDM symbol index)-frequency index(sub-carrier index) space for accommodating data coded bits suitable forthe practice of the principles of another embodiment of the presentinvention;

FIG. 18 shows an implementation of rate matching and bit collection on acode block basis suitable for the practice of the principles of anotherembodiment of the present invention;

FIG. 19 shows an example of spreading the coded bits of a code blockover time, frequency, and spatial domains suitable for the practice ofthe principles of another embodiment of the present invention;

FIG. 20 shows an example of writing coded bits into resources withdifferent layers and different modulation orders suitable for thepractice of the principles of another embodiment of the presentinvention;

FIG. 21 illustrates channel interleaver with different modulation orderon resources suitable for the practice of the principles of anotherembodiment of the present invention;

FIG. 22 shows an example of spread coded bits on resources withdifferent spatial dimensions suitable for the practice of the principlesof another embodiment of the present invention;

FIG. 23 shows a general method of obtaining the preferred permutationpattern for 64-QAM suitable for the practice of the principles ofanother embodiment of the present invention;

FIG. 24 illustrates an iterative receiver for decoding multiple codeblocks multiplexed within same modulation symbols suitable for thepractice of the principles of another embodiment of the presentinvention;

FIG. 25 shows an example of a reduced constellation which improves thedetection performance of the transmission suitable for the practice ofthe principles of another embodiment of the present invention;

FIG. 26 illustrates an alternative iterative receiver for decodingmultiple code clocks multiplexed in same modulation symbols suitable forthe practice of the principles of another embodiment of the presentinvention;

FIG. 27 shows the downlink subframe structure in a 3rd GenerationPartnership Project (3GPP) Long Term Evolution (LTE) system;

FIG. 28 shows another example of the configuration of grouping multipleOFDM symbols in a subframe suitable for the practice of the principlesof one embodiment of the present invention;

FIG. 28( a) is a flow chart illustrating a method of transmitting datasignals by separating resource elements having coded bits suitable forthe practice of the principles of one embodiment of the presentinvention.

FIG. 28 (b) is a flow chart illustrating a method of receiving anddecoding grouped resource elements having coded bits at a receiver.

FIG. 29 shows another example of the configuration of grouping multipleOFDM symbols in a subframe suitable for the practice of the principlesof one embodiment of the present invention;

FIG. 30 shows another example of the configuration of grouping multipleOFDM symbols in a subframe suitable for the practice of the principlesof one embodiment of the present invention;

FIG. 31 shows examples of parallel processing for successiveinterference cancellation either with or without grouping cyclic delaydiversity (CRC) suitable for the practice of the principles of thepresent invention; and

FIG. 32 shows an example of the configuration of grouping code bocksuitable for the practice of the principles of another embodiment of thepresent invention.

DETAILED DESCRIPTION OF THE INVENTION

In the detailed description of this invention, following terms will befrequently used and the definition of each term is provided.

A subpacket is a portion of an encoded packet, and is a subset of thetotal coded bits.

Data bits are a stream of information bits which are encoded to generatecoded bits.

Interlace refers to a subset of transmission slots or sub-frames.

The synchronous hybrid automatic repeat request (S-HARQ) is a techniqueemployed by the current high rate packet data (HRPD) standard, whichestablishes a set of four time-division interlaced transmission channelsused for the concurrent transmission of four different sets of data.These interlaced transmission channels are sometimes referred to as“HARQ interlaces”.

A transmission slot is an allocated predetermined number of consecutiveclock cycles. A number of these transmission slots form a transmissionframe.

Space-time coding (STC) is a method employed to improve the reliabilityof data transmission in wireless communication systems using multipletransmit antennas. STCs rely on transmitting multiple, redundant copiesof a data stream to the receiver so that at least some of the datastream copies may survive in the physical path between transmission andreception in a good state to allow reliable decoding.

Transmit diversity method is a method that one data bit is transmittedvia different independent channels.

Receiver diversity method is a method that one data bit is received viadifferent independent channels.

Channel Quality Indicator (CQI) is a measurement of the communicationquality of wireless channels. CQI (channel quality indicator) can be avalue (or values) representing a measure of channel quality for a givenchannel.

Redundancy Version Parameter indicates which redundancy version of thedata is sent.

Channel interleaver sends data interleaved via different channels inorder that deep fade or collision at some channels does not void thetransmission.

Resource block is a block of time and frequency resource elements thatcarry signals to be transmitted by the transmitter and to be received bythe receiver.

Methods and apparatus to enable fast decoding of transmissions withmultiple code blocks constructed according to the present invention willbe described in details with reference to the accompanying drawings.Like reference numerals designate like elements throughout thespecification.

Also, several acronyms frequently used in this invention are listed asbelow with their own full names.

SC-FDMA: Single Carrier Frequency Division Multiple Access

CP: cyclic prefix

FFT: Fast Fourier Transform

OFDM: Orthogonal Frequency Division Multiplexing

ICI: Inter-Carrier-Interference

3GPP: 3rd Generation Partnership Project

LTE: Long Term Evolution

HARQ: Hybrid Automatic Repeat reQuestion

MIMO: Multiple Input Multiple Output

QPSK: Quadrature phase shift keying

16-QAM: 16 Quadrature amplitude modulation

64-QAM: 64 Quadrature amplitude modulation

IFFT: Inverse Fast Fourier Transform

CW: codeword

Code block: a block of data bits or the block of coded bits generated byencoding the block of data bits

FIG. 1 shows an Orthogonal Frequency Division Multiplexing (OFDM)transceiver chain having transmitter chain and receiver chain.

Orthogonal Frequency Division Multiplexing (OFDM) is a technology tomultiplex data in frequency domain. Modulation symbols are carried onfrequency sub-carriers. A sample of Orthogonal Frequency DivisionMultiplexing (OFDM) transceiver chain is shown in FIG. 1. At atransmitter chain 100, control signals or data signals are modulated bya modulator 101 and the modulated signals are serial-to-parallelconverted by a serial-to-parallel convertor 112. An Inverse Fast FourierTransform (IFFT) unit 114 is used to transfer the modulated signal ordata from frequency domain to time domain, and the modulated signalstransferred to time domain is parallel-to-serial converted by aparallel-to-serial convertor 116. A cyclic prefix (CP) or zero prefix(ZP) is added to each OFDM symbol at a CP insertion stage 118 to avoidor mitigate the impact due to multipath fading at a multipath fadingchannel 122. Signals from cyclic prefix (CP) insertion stage 118 aretransmitted to transmitter front end processing unit 120, for example,transmit antennas (not shown on FIG. 1). At a receiver chain 140,assuming perfect time and frequency synchronization are achieved,signals received by receiver front end processing unit 124, for example,receive antennas (not shown on FIG. 1), are processed at a cyclic prefix(CP) removal stage 126 in which removes the cyclic prefix (CP) of thereceived signal. Signal processed at cyclic prefix (CP) removal stage126 is further serial-to-parallel converted by a serial-to-parallelconvertor 128. A Fast Fourier Transform (FFT) unit 130 transfers thereceived signals from time domain to frequency domain for furtherprocessings, such as being parallel-to-serial converted by aparallel-to-serial convertor 132 and being demodulated by a de-modulatorTherefore, the signals transmitted by transmitter chain 100 are receivedby receiver chain

FIG. 2 illustrates the orthogonality of Orthogonal Frequency DivisionMultiplexing (OFDM) theory.

Because each OFDM symbol has finite duration in time domain, thesub-carriers overlap with each other in frequency domain. For example,as shown in FIG. 2, sub-carrier0 10, sub-carrier1 11 and sub-carrier2 12overlap with each other in frequency domain. sub-carrier0 10,sub-carrier1 11 and sub-carrier2 12 have almost identical or similarwave shapes. These three sub-carriers are mathematically perpendicularto each other, in other words, the inner products of any two of thesub-carriers are zero. The orthogonality of Orthogonal FrequencyDivision Multiplexing (OFDM) theory, therefore, is maintained at thesampling frequency assuming the transmitter and receiver has perfectfrequency synchronization. In the case of frequency offset due toimperfect frequency synchronization or high mobility, the orthogonalityof the sub-carriers at sampling frequencies is destroyed, resulting inInter-Carrier-Interference (ICI).

FIG. 3 a is an illustration of a transmitted OFDM symbol in time domain,and FIG. 73 b is an illustration of the received OFDM symbols in timedomain.

As shown in FIG. 3 a, A multipath fading channel may be approximated asimpulse response channel in time domain, and may be presented as afrequency selective channel in frequency domain. Because of multipathfading channel 122 in the Orthogonal Frequency 1. Division Multiplexing(OFDM) transceiver chain as shown in FIG. 1, a CP portion inserted toone received symbol is often corrupted by the previous OFDM symbol.Transmit signal 20 has continuously transmitted OFDM symbols (i.e. OFDMSymbol 1, OFDM Symbol 2, . . . ), and cylic prefix (CP) portions (i.e.CP1 and CP2) are located between any of two OFDM Symbols. Aftertransmitted through multipath fading channel 122, receive signal 27 hascontinuously CP inserted OFDM symbols (i.e. Rx OFDM Symbol1 28, Rx OFDMSymbol2 29, . . . ). Rx OFDM Symbol1 28 and Rx OFDM Symbol2 29 arecorrupted by their own CP, respectively. For example, CP3 corrupts intoRx OFDM Symbol1 28. When the length of cylic prefix (CP) is sufficientlylong, the received OFDM symbols without cylic prefix (CP) portion,however, should only contain their own signal convoluted by themultipath fading channel. In general, a FFT process by FFT unit 130 asshown in FIG. 1 is taken at the receiver side to allow furtherprocessing in frequency domain. The advantage of Orthogonal FrequencyDivision Multiplexing (OFDM) over other transmission schemes is therobustness to multipath fading. The multipath fading in time domaintranslates into frequency selective fading in frequency domain. With thecyclic prefix or zero prefix inserted, the inter-symbol-interferencebetween adjacent OFDM symbols are avoided or largely alleviated.Moreover, because each modulation symbol is carried over a narrowbandwidth, each modulation symbol experiences a single path fading.Simple equalization scheme may be used to combat frequency selectivefading.

Single Carrier Frequency Division Multiple Access (SC-FDMA), whichutilizes single carrier modulation and frequency domain equalization isa technique that has similar performance and complexity as those of anOrthognal Frequency Division Multiplexing Access (OFDMA) system. Oneadvantage of Single Carrier Frequency Division Multiple Access (SC-FDMA)is that the Single Carrier Frequency Division Multiple Access (SC-FDMA)signal has lower peak-to-average power ratio (PAPR) because SingleCarrier Frequency Division Multiple Access (SC-FDMA) has an inherentsingle carrier structure. Low PAPR normally results in high efficiencyof power amplifier, which is particularly important for mobile stationsin uplink transmission. Single Carrier Frequency Division MultipleAccess (SC-FDMA) is selected as the uplink multiple access scheme in3GPP Long Term Evolution (LTE).

FIG. 4 shows an example of a transceiver chain for Single CarrierFrequency Division Multiple Access (SC-FDMA).

An example of the transceiver chain for Single Carrier FrequencyDivision Multiple Access (SC-FDMA) is shown in FIG. 4. At transmitterchain 200, time-domain data or control data is modulated by a modulator201, and the modulated data is serial-to-parallel converted by aserial-to-parallel convertor 212. A Discrete Fourier Transform (DFT)unit 213 processes the converted data by a discrete fourier transformprocess. To ensure low PAPR, the transformed data is then mapped to aset of contiguous sub-carriers at a subcarrier mapping stage 211. Thenan IFFT unit 214 transforms the signal back to time domain, and IFFTunit normally has a larger IFFT size than that of DFT unit 213. Aparallel-to-serial convertor 216 parallel-to-serial converts thereceived data. Cyclic prefix (CP) is added at CP insertion stage 228before data is transmitted and processed by a transmission front endprocessing unit 220. Front end processing unit 220 has an amplificationstage enabling wireless transmission of the plurality of groups of theencoded data bits in a predetermined sequence via a plurality oftransmitting antennas. The processed signal with a cyclic prefix addedis often referred to as a Single Carrier Frequency Division MultipleAccess (SC-FDMA) block. After the processed signal passes through acommunication channel, e.g., a multipath fading channel 222 in awireless communication system, a receiver chain 240 performs a receiverfront end processing at a receiver front end processing unit 224,removes the cylic prefix (CP) by a CP remover 226, serial-to-parallelconverts the data by a serial-to-parallel convertor 228, transforms thedata by a FFT unit 230 and demaps the data at a subcarrierdemapping/equalization unit 231 in frequency domain. Inverse DiscreteFourier Transform (IDFT) unit 233 processes data after the equalizedsignal is demapped in frequency domain. The output of IDFT unit 235 isfurther processed by a parallel-to-serial convertor 232 and ademodulator 236.

FIG. 5 is an illustration of Hybrid Automatic Repeat reQuestion (HARQ)operation.

Hybrid Automatic Repeat reQuestion (HARQ) is widely used incommunication systems to combat decoding failure and improve thereliability of data transmission. A HARQ operation is shown in FIG. 5. Adata packet is coded by using an encoder 311 with a certain kind ofForward Error Correction (FEC) scheme. The data packet is processed by asubpacket generator 312 and a set of subpackets are generated. Asubpacket, for example, a subpacket k may only contain a portion of thecoded bits. If the transmission by a transceiver 300 for subpacket kfails, as indicated by a NAK negative acknowledgement provided by afeedback acknowledgement channel 314, a retransmission subpacket,subpacket k+1, is provided to retransmit this data packet. If subpakcetk+1 is successfully transceived, an ACK acknowledgement is provided byfeedback acknowledgement channel 314. The retransmission subpackets maycontain different coded bits from previous subpackets. The receiver maysoftly combine or jointly decode all the received subpackets by adecoder 313 to improve the chance of decoding. Normally, a maximumnumber of transmissions is configured in consideration of bothreliability, packet delay, and implementation complexity.

N-channel synchronous Hybrid Automatic Repeat reQuestion (HARQ) is oftenused in wireless communication systems because of the simplicity. Forexample, synchronous Hybrid Automatic Repeat reQuestion (HARQ) has beenaccepted as the Hybrid Automatic Repeat reQuestion (HARQ) scheme forlong term evolution (LTE) uplink in 3GPP.

FIG. 6 shows an example of a four-channel synchronous Hybrid AutomaticRepeat reQuestion (HARQ).

Because of the fixed timing relationship between subsequenttransmissions, the transmission slots in an individual HARQ channelexhibits an interlace structure. For example, interlace 0 includes slot0, 4, 8, . . . , 4 k, . . . ; interlace 1 includes slot 1, 5, 9, . . . ,4 k+1, . . . ; interlace 2 includes slot 2, 6, 10, . . . , 4 k+2, . . .; interlace 3 includes slot 3, 7, 11, . . . 4 k+3. . . . A packet istransmitted in slot 0. After correctly decoding the packet, the receiversends back an ACK acknowledgement to the transmitter. The transmitterthen starts transmitting a new packet at the next slot in thisinterlace, i.e., slot 4. The first subpacket of the new packettransmitted in slot 4, however, is not properly received. After thetransmitter receives a NAK negative acknowledgement from the receiver,the transmitter transmits another sub-packet of the same packet at thenext slot in interlace 0, i.e., slot 8. Interlaces 1-3 act in similarways as interlace 0. Sometimes the receiver may have difficulty indetecting the packet boundary, i.e., whether a subpacket is the firstsub-packet of a new packet or a retransmission sub-packet. To alleviatethis problem, a new packet indicator may be transmitted in a controlchannel that carries transmission format information for the packet.Sometimes, a more elaborated version of HARQ channel information, suchas sub-packet ID, and/or HARQ channel ID, may be provided to help thereceiver detect and decode the packet.

Multiple antennas communication systems, which are often referred to asMultiple Input Multiple Output (MIMO), are widely used in wirelesscommunication to improve system performance. In a MIMO system, thetransmitter has multiple antennas capable of transmitting independentsignals and the receiver is equipped with multiple receive antennas.MIMO systems degenerates to Single Input Multiple Output (SIMO) if thereis only one transmit antenna or if there is only one stream of datatransmitted. MIMO systems degenerates to Multiple Input Signal Output(MISO) if there is only one receive antenna. MIMO systems degenerates toSingle Input Single Output (SISO) if there is only one transmit antennaand one receive antenna. MIMO technology may significant increasethroughput and range of the system without any increase in bandwidth oroverall transmit power. In general, MIMO technology increases thespectral efficiency of a wireless communication system by exploiting theadditional dimension of freedom in the space domain due to multipleantennas. There are many categories of MIMO technologies. For example,spatial multiplexing schemes increase the transmission rate by allowingmultiple data streaming transmitted over multiple antennas. Transmitdiversity methods such as space-time coding take advantage of spatialdiversity due to multiple transmit antennas. Receiver diversity methodsutilize the spatial diversity due to multiple receive antennas.Beamforming technologies improve received signal gain and reducinginterference to other users. Spatial Division Multiple Access (SDMA)allows signal streams from or to multiple users to be transmitted overthe same time-frequency resources. The receivers can separate themultiple data streams by the spatial signature of these data streams.Note that these MIMO transmission techniques are not mutually exclusive.In fact, multiple MIMO schemes may be used in an advanced wirelesssystems.

When the channel is favorable, e.g., the mobile speed is low, aclosed-loop MIMO scheme may be used to improve system performance. In aclosed-loop MIMO systems, receivers provide the feedback of channelcondition and/or preferred transmitter MIMO processing schemes. Thetransmitters may utilize this feedback information, together with otherconsiderations such as scheduling priority, data and resourceavailability, to jointly optimize the transmission scheme.

A popular closed-loop MIMO scheme is called MIMO precoding. During apreceding process, the data streams to be transmitted are precoded, i.e.pre-multiplied by a matrix, before being passed on to the multipletransmit antennas.

FIG. 7 shows a Multiple Input Multiple Output (MIMO) system.

As shown in FIG. 7, a transmitter 401 has a number of Nt transmitantennas 411 and a receiver 402 has a number of Nr receive antenna 421.Data streams 1-Ns are transceived by this MIMO system. A matrix H isdenoted as a transceive channel between transmit antennas 411 andreceive antennas 421, and channel H is a Nt by Nr matrix. If transmitter401 has the knowledge of channel matrix H, transmitter 401 can choosethe most advantageous transmission scheme based on channel matrix H. Forexample, when maximizing throughput is the goal of the transmissionsystem, a preceding matrix may be chosen to be the right singluar matrixof channel matrix H if the knowledge of H is available at transmitter401. Therefore, the effective channel for the multiple data streams atthe receiver side can be diagonalized, interference between the multipledata streams may be eliminated. The overhead required to feedback theexact value of channel H, however, is often prohibitive.

FIG. 8 shows a precoding process as used in a closed-loop MIMO system.

As shown in FIG. 8, Data streams 1-Ns are processed by a process stage510 where a scheduling process, a power and rate adapation process, aprecoding codebook and preceding vector selection process, astream-to-layer mapping process and some other related processes areexecuted. Mapped data streams are transmitted via Layers 1-N_(L) to aprecoding stage 509, i.e. transmitter MINO processing stage. Theprecoded data is further transmitted to transmit antennas 1-Nt. Areceiver 512 receives and restores data streams 1-Nr at a receiver MINOprocessing stage 508. In order to reduce a feedback overhead, multipleprecoding matrices are defined at transmitter 511 to quantize the spaceof the possible values that channel matrix H could substantiate. Withthe space quantization, receiver 512 feeds back the preferred precodingscheme, normally in the form of the index of the preferred precedingmatrix, the transmission rank, and the indices of the preferredpreceding vectors. Receiver 512 may also feed back an associated ChannelQuality Indication (CQI) values for the preferred precoding scheme.

Another perspective of a MIMO system is whether the multiple datastreams to be transmitted are encoded separately or together. If all ofthe transmission layers are encoded together, this MIMO system is calledSingle CodeWord (SCW) MIMO system, otherwise is called a MultipleCodeWord (MCW) MIMO system. In the long term evolution (LTE) downlinksystem, when Single User MIMO (SU-MIMO) is used, up to two MIMOcodewords can be transmitted to a single User Equipment (UE). In thecase that two MIMO codewords are transmitted to a User Equipment (UE),the UE needs to acknowledge these two codewords separately. Another MIMOtechnique is called Spatial Division Multiple Access (SDMA), which isalso referred to as Multi-User MIMO (MU-MIMO) sometimes. In SDMA,multiple data streams are encoded separately and transmitted todifferent intended receivers on the same time-frequency resources. Byusing different spatial signatures, e.g., antennas, virtual antennas, orprecoding vectors, the receivers will be able to distinguish themultiple data streams. Moreover, by scheduling a proper group ofreceivers and choosing the proper spatial signature for each data streambased on channel state information, the signal of interest can beenhanced for the receiver of interest while the other signals can beenhanced for the other corresponding receivers at the same time.Therefore the system capacity may be improved. Both single user MIMO(SU-MIMO) and multi-user MIMO (MU-MIMO) are adopted in the downlink oflong term evolution (LTE). MU-MIMO is also adopted in the uplink of longterm evolution (LTE), SU-MIMO for long term evolution (LTE) uplink,however, is still under discussion.

In a long term evolution (LTE) system, when a transport block is large,the transport block is segmented into multiple code blocks so thatmultiple coded packets can be generated, which is advantageous becauseof benefits such as enabling parallel processing and pipelineimplementation and flexible trade-off between power consumption andhardware complexity.

As an example, the encoding process of the High Speed Data SharedChannel (HS-DSCH) in a High Speed Downlink Packet Access (HSDPA) systemis illustrated in FIG. 9.

As shown in FIG. 9, data bits a_(im1), a_(im2), a_(im3) . . . a_(imA)are processed at a cyclic redundancy check (CRC) stage 611 and aretransformed to data bits b_(im1), b_(im2), b_(im3) . . . b_(imB). CRCattached data bits are bit scrambled at a bit scrambling stage 612, andare transformed to data bits d_(im1), d_(im2), d_(im3) . . . d_(imB).Scrambled data bits are segmented at code block segmentation stage 613and are formed to code blocks o_(ir1), o_(ir2), o_(ir3) . . . o_(irK).Code blocks then are coded at a channel coding stage 614 and becomecoded code blocks c_(ir1), c_(ir2), c_(ir3) . . . c_(irE). These codedcode blocks are processed at a physical layer Hybrid-ARQ functionalitystage 615. The resulting data bits are again segmented at a physicalchannel segmentation stage 616. The Hybrid-ARQ functionality matches thenumber of bits w₁, w₂, w₃ . . . w_(R) at the coded bits to the totalnumber of bits of HS-DSCH physical channel. The resulting channelsegmented data bits u_(p,1), u_(p,2), u_(p,3) . . . u_(p,U) areinterleaved by a HS-DSCH interleaving stage 617. Interleaved data bitsv_(p,1), v_(p,2), v_(p,3) . . . v_(p,U) are then re-arranged at aconstellation re-arrangement stage 618 and re-arranged bits r_(p,1),r_(p,2), r_(p,3) . . . r_(p,U) are further mapped at a physical channelmapping stage 619. The resulting mapped bits are finally output tophysical channel #1, physical channel #2 . . . physical channel #P. Incurrent HS-DSCH design, only one 24-bit cyclic redundancy check (CRC) isgenerated for the whole transport block for the purpose of errordetection for that block. If multiple code blocks are generated andtransmitted in one Transmission Time Interval (TTI), the receiver maycorrectly decode some of the code blocks but not the others. In thatcase, the receiver has to feedback a is NAK negative acknowledgement tothe transmitter because the CRC for the transport block will not check.

The hybrid ARQ functionality matches the number of bits at the output ofthe channel coder (i.e. channel coding stage 614) to the total number ofbits of the HS-PDSCH set to which the HS-DSCH is mapped. The hybrid ARQfunctionality is controlled by redundancy version (RV) parameters. Theexact set of bits at the output of the hybrid ARQ functionality dependson the number of input bits, the number of output bits, and the RVparameters.

The hybrid ARQ functionality has two rate-matching stages and a virtualbuffer as shown in FIG. 10.

A stream of data bits N^(TTI) from resource C is separated intosystematic bits, parity1 bits and parity2 bits by a bit separator 610.These three groups of bits are processed differently at a first ratematching stage 611. First rate matching stage 611 matches the number ofinput bits to a virtual IR buffer 613, and the information about buffer613 is provided by higher layers. Systematic bits are directly providedto buffer 613, parity1 bits are processed by rate matcher RM_P1_1 andparity2 bits are processed by rate matcher RM_P2_1. Outputs of buffer613 are provided to a second rate matching stage 615. Second ratematching stage 615 matches the number of bits after first rate matchingstage 611 to the number of physical channel bits available in theHS-PDSCH set in the Transmission Time Interval (TTI). Output N_(sys) isprovided to rate matcher RM_S of second rate matching stage 615, outputN_(p1) is provided to rate matcher RM_P1_2 of second rate matching stage615, and output N_(p2) is provided to rate matcher RM_P2_2 of secondrate matching stage 615. Outputs N_(sys), N_(p1), and N_(p2) areprovided to a bit collection stage 617. Therefore, a resulting data bitsstream N_(data) is provided to terminal W. Note that, if the number ofinput bits does not exceed buffering capability virtual IR buffer 613,first rate-matching stage 611 is transparent.

Different modulation schemes, such as Quadrature phase shift keying(QPSK), binary phase shift keying (BPSK), 8 Phase-shift keying (8-PSK),16 Quadrature amplitude modulation (16-QAM), or 64 Quadrature amplitudemodulation (64-QAM) may be used for an adaptive modulation andincreasing the spectral efficiency of modulation. In case of 16-QAMmodulation, quadruples of bits, b0b1b2b3, are mapped to complex-valuedmodulation symbols x=I+jQ. One implementation of 16-QAM is illustratedin Table 1.

TABLE 1 16-QAM modulation mapping b₀b₁b₂b₃ I Q 0000  1/{square root over(10)}  1/{square root over (10)} 0001  1/{square root over (10)} 3/{square root over (10)} 0010  3/{square root over (10)}  1/{squareroot over (10)} 0011  3/{square root over (10)}  3/{square root over(10)} 0100  1/{square root over (10)} −1/{square root over (10)} 0101 1/{square root over (10)} −3/{square root over (10)} 0110  3/{squareroot over (10)} −1/{square root over (10)} 0111  3/{square root over(10)} −3/{square root over (10)} 1000 −1/{square root over (10)} 1/{square root over (10)} 1001 −1/{square root over (10)}  3/{squareroot over (10)} 1010 −3/{square root over (10)}  1/{square root over(10)} 1011 −3/{square root over (10)}  3/{square root over (10)} 1100−1/{square root over (10)} −1/{square root over (10)} 1101 −1/{squareroot over (10)} −3/{square root over (10)} 1110 −3/{square root over(10)} −1/{square root over (10)} 1111 −3/{square root over (10)}−3/{square root over (10)}

The constellation of the 16-QAM modulation in Table 1 is shown in FIG.11. FIG. 11 shows a two dimensional coordinate which shows oneillustration of 16-QAM constellation diagram. A constellation diagram isa representation of a signal modulated by a digital modulation scheme.The constellation diagram displays signals on a two-dimensionalcoordinate diagram in a complex plane at symbol sampling instants. Theconstellation diagram represents the possible symbols selected by agiven modulation scheme as dots in the complex plane. Each dot on FIG.11 illustrates a corresponding symbol of b0b1b2b3 on the I-Q complexplan when I and Q have predetermined values as show in Table 1. Thisconstellation provides different protection levels on the four bits(i.e. bit b0, b1, b2 and b3). As shown in FIG. 11, the protection levelon bits b0 and b1 are the same, the protection level on bits b2 and b3are the same. The protection level on b0 and b1, however, are higherthan the protection level on bits b2 and b3.

In case of 64QAM modulation, sextuplets of bits, b0b1b2b3b4b5, aremapped to complex-valued modulation symbols x=I+jQ. One implementationof 64-QAM is shown in Table 2. The constellation of the 64-QAMmodulation in Table 2 is shown in FIG. 11. FIG. 12, shows a twodimensional coordinate which shows one illustration of 64-QAMconstellation diagram. This constellation provides different protectionlevels on the six bits. Similar to FIG. 11, each dot on FIG. 12illustrates a corresponding symbol of b0b1b2b3b4b5 on the I-Q complexplan when I and Q have predetermined values as show in Table 2. Theprotection level on bits b0 and b1 are the same, the protection on bitsb2 and b3 are the same, and the protection level on bits b4 and b5 arethe same. The protection levels on bits b0 and b1, however, are strongerthan the protection levels on bits b2 and b3 that are stronger than theprotection on bits b4 and b5. For convenience, the index of a bit isdefined in a modulation symbol as the modulation position of that bit.

TABLE 2 64-QAM modulation mapping b₀b₁b₂b₃b₄b₅ I Q 000000  3/{squareroot over (42)}  3/{square root over (42)} 000001  3/{square root over(42)}  1/{square root over (42)} 000010  1/{square root over (42)} 3/{square root over (42)} 000011  1/{square root over (42)}  1/{squareroot over (42)} 000100  3/{square root over (42)}  5/{square root over(42)} 000101  3/{square root over (42)}  7/{square root over (42)}000110  1/{square root over (42)}  5/{square root over (42)} 000111 1/{square root over (42)}  7/{square root over (42)} 001000  5/{squareroot over (42)}  3/{square root over (42)} 001001  5/{square root over(42)}  1/{square root over (42)} 001010  7/{square root over (42)} 3/{square root over (42)} 001011  7/{square root over (42)}  1/{squareroot over (42)} 001100  5/{square root over (42)}  5/{square root over(42)} 001101  5/{square root over (42)}  7/{square root over (42)}001110  7/{square root over (42)}  5/{square root over (42)} 001111 7/{square root over (42)}  7/{square root over (42)} 010000  3/{squareroot over (42)} −3/{square root over (42)} 010001  3/{square root over(42)} −1/{square root over (42)} 010010  1/{square root over (42)}−3/{square root over (42)} 010011  1/{square root over (42)} −1/{squareroot over (42)} 010100  3/{square root over (42)} −5/{square root over(42)} 010101  3/{square root over (42)} −7/{square root over (42)}010110  1/{square root over (42)} −5/{square root over (42)} 010111 1/{square root over (42)} −7/{square root over (42)} 011000  5/{squareroot over (42)} −3/{square root over (42)} 011001  5/{square root over(42)} −1/{square root over (42)} 011010  7/{square root over (42)}−3/{square root over (42)} 011011  7/{square root over (42)} −1/{squareroot over (42)} 011100  5/{square root over (42)} −5/{square root over(42)} 011101  5/{square root over (42)} −7/{square root over (42)}011110  7/{square root over (42)} −5/{square root over (42)} 011111 7/{square root over (42)} −7/{square root over (42)} 100000 −3/{squareroot over (42)}  3/{square root over (42)} 100001 −3/{square root over(42)}  1/{square root over (42)} 100010 −1/{square root over (42)} 3/{square root over (42)} 100011 −1/{square root over (42)}  1/{squareroot over (42)} 100100 −3/{square root over (42)}  5/{square root over(42)} 100101 −3/{square root over (42)}  7/{square root over (42)}100110 −1/{square root over (42)}  5/{square root over (42)} 100111−1/{square root over (42)}  7/{square root over (42)} 101000 −5/{squareroot over (42)}  3/{square root over (42)} 101001 −5/{square root over(42)}  1/{square root over (42)} 101010 −7/{square root over (42)} 3/{square root over (42)} 101011 −7/{square root over (42)}  1/{squareroot over (42)} 101100 −5/{square root over (42)}  5/{square root over(42)} 101101 −5/{square root over (42)}  7/{square root over (42)}101110 −7/{square root over (42)}  5/{square root over (42)} 101111−7/{square root over (42)}  7/{square root over (42)} 110000 −3/{squareroot over (42)} −3/{square root over (42)} 110001 −3/{square root over(42)} −1/{square root over (42)} 110010 −1/{square root over (42)}−3/{square root over (42)} 110011 −1/{square root over (42)} −1/{squareroot over (42)} 110100 −3/{square root over (42)} −5/{square root over(42)} 110101 −3/{square root over (42)} −7/{square root over (42)}110110 −1/{square root over (42)} −5/{square root over (42)} 110111−1/{square root over (42)} −7/{square root over (42)} 111000 −5/{squareroot over (42)} −3/{square root over (42)} 111001 −5/{square root over(42)} −1/{square root over (42)} 111010 −7/{square root over (42)}−3/{square root over (42)} 111011 −7/{square root over (42)} −1/{squareroot over (42)} 111100 −5/{square root over (42)} −5/{square root over(42)} 111101 −5/{square root over (42)} −7/{square root over (42)}111110 −7/{square root over (42)} −5/{square root over (42)} 111111−7/{square root over (42)} −7/{square root over (42)}For example, the modulation position of b0 in a 64-QAM is 0, themodulation position of b1 in a 64-QAM is 1. Therefore, for the given64-QAM constellation, the first and second modulation positions, i.e.,b0 and b1, have the strongest protection; the third and fourthmodulation positions, i.e., b2 and b3, have weaker protection levels;the fifth and the sixth modulation positions, i.e., b4 and b5, have theweakest protection level.

In this invention, methods and apparatus are provided to improve theperformance of transmissions with information bits or parity bits frommultiple coded packets.

Aspects, features, and advantages of the invention are readily apparentfrom the following detailed description, simply by illustrating a numberof particular embodiments and implementations, including the best modecontemplated for carrying out the invention. The present invention isalso capable of other and different embodiments, and several details ofthe present invention may be modified into various obvious respects, allwithout departing from the spirit and scope of the invention.Accordingly, the drawings and description are to be regarded asillustrative in nature, and not as restrictive. The present invention isillustrated by way of example, and not by way of limitation, in thefigures of the accompanying drawings.

In the following illustrations, downlink data channel in long termevolution (LTE) systems are used as an example. However, the techniqueillustrated here can certainly be used in uplink data channel in longterm evolution (LTE) systems, control channels in either downlink oruplink in long term evolution (LTE) systems, and other data, control, orother channels in other systems whenever applicable.

In the present invention, an improved design of channel interleaver andreceiver is provided and a separate coding method of multiple codeblocks is taken into account to improve the performance. When multiplecode blocks are transmitted, the performance of the transmission isdictated by the code block that has the worst performance. The idea isto carefully design the channel interleaver, including the mapping fromcoded bits of different code blocks to modulation symbols, and themapping from modulation symbols to time, frequency, and spatialresources, to make sure each code block get roughly the same level ofprotection. On the receiver side, when some code blocks are receivedcorrectly and some are not, the signal of the successfully decoded codeblocks may be reconstructed and cancelled from the received signal.After the cancellation, the receiver may attempt to re-decode the othercode blocks. Because the interference with other code blocks that arenot yet successfully decoded may be greatly reduced, the probabilitythat the receiver will be able to decode the other code blocks can besignificantly increased. In the case of hybrid ARQ (HARQ), if thereceiver is not able to decode one of the code blocks, the receiver willfeedback NAK for the whole transport block, assuming there is only oneACK channel. Because the Node B has no knowledge which code block issuccessfully decoded by the UE and which is not, the Node B willretransmit as if the whole transport block including all code blocks areNAKed. In that case, UE should be able to utilize the knowledge aboutthose successfully decoded code blocks to help decoding those codeblocks that have not been successfully decoded. The channel interleaverdesign proposed in this invention facilitates that operation. Preferredembodiments of the receiver operation are also disclosed.

In one embodiment of the invention, a CRC is added to each code block toenable error detection for each code block. FIG. 13 shows an example ofthe attachment of code block CRC. Comparing to FIG. 9, after thetransport block CRC attachment, the bit scrambling, and the code blocksegmentation, an additional step of attaching code block CRC to at leastone of the code blocks follows immediately after the step of code clocksegmentation as shown in FIG. 13. The transport block is segmented intoone or multiple code blocks. If there is only one code block in thetransport block, the code block CRC may not be necessary. If there ismore than one code block in the transport block, the attachment of codeblock CRC becomes important. As an example, the transport block CRC forHS-DSCH in HSDPA is 24-bit, which provides very low detection error(about 2⁻²⁴≈6×10⁻⁸). One purpose of attaching CRC to each code block isto provide sufficient code block error detection so that the receivercan cancel the signals of those code blocks that are correctly decoded.A CRC detection error of ˜10⁻² may be sufficient for this operation.Note that an 8-bit CRC can provide a detection error rate of about4×10⁻³. In this case, an 8-bit CRC may be used for code block CRC forcode block error detection and cancellation, while a 24-bit CRC may beused for transport block error detection. By doing so, the CRC overheadis minimized while providing means to facilitate cancellation ofsuccessfully decoded code blocks. Obviously, the CRC overhead can befurther reduced by only attaching one code block CRC for multiple codeblocks.

In the present invention, a number of steps are provided to be appliedin channel interleaver design. Note that not all of these steps need tobe incorporated in order to use this invention. In other words, thepresent invention covers the interleavers and interleaving methods thatuse at least one of the steps illustrated in this invention. Note thatthe constellation re-arrangement for 16-QAM as shown in FIG. 13 may notbe necessary with the proposed channel interleaver design.

Turning now to FIG. 14, FIG. 15 and FIG. 16.

FIG. 14 illustrates channel interleaver for Orthogonal FrequencyDivision Multiplexing (OFDM) systems suitable for the practice of theprinciples of one embodiment of the present invention. FIG. 15illustrates a resource element map for a data transmission suitable forthe practice of the principles of one embodiment of the presentinvention. FIG. 16 shows a re-arrangement of coded bits by code blocksafter rate matching suitable for the practice of the principles of oneembodiment of the present invention. As shown in FIG. 14, the resourceallocated to a data transmission is N OFDM symbols in time, and Msub-carriers in frequency. Each sub-carrier of one OFDM symbol is calledone resource element. Resource elements include data resource elementsand non-data resource elements. Each resource element carries amodulation symbol, which in turn carries multiple coded bits. Forexample, 2 bits can be carried in one QPSK modulation symbol; 4 bits canbe carried in one 16-QAM modulation symbol, etc. The number of bitscarried in each modulation symbol is denoted as a modulation order L,each modulation position in a modulation symbol is represented by onemodulation position index as shown in FIG. 14. One preferred embodimentof the channel interleaver design consists at least one of the followingoperations.

Step 1

First, for each code block, symbols S, P₁, P₂, are, respectively, thesystematic bits, parity bits from encoder 1 of a turbo encoder, andparity bits from encoder 2 of the turbo encoder. A turbo encoder isformed by parallel concatenation of two recursive systematicconvolutional (RSC) encoders separated by an interleaver. In oneembodiment of the present invention, the coded bits after the secondrate matching are re-arranged based on code blocks. As illustrated inFIG. 16, there are a number of N_(cb) code blocks in transmission forthis transport block. Comparing to FIG. 10, a stage 913 called bitsre-arrangement by code blocks immediately follows bit collection stage617. In stage 913, the systematic bits, parity 1 bits, and parity 2 bitsof code block 1 are collected together and arranged in the order of S,P₁, P₂. The number of systematic bits, parity 1 bits, and parity 2 bitsof the i-th code block are denoted by N_(t,i,sys), N_(t,i,p1),N_(t,i,p2), respectively. The re-arranged bits then enter a stage 915,which is called channel interleaver stage. N_(t,sys), N_(tp1), N_(t,p2)may be presented as following equations respectively.

$\begin{matrix}{N_{t,{sys}} = {\sum\limits_{i = 1}^{Ncb}N_{t,i,{sys}}}} & (1) \\{N_{t,{p\; 1}} = {\sum\limits_{i = 1}^{Ncb}N_{t,i,{p\; 1}}}} & (2) \\{N_{t,{p\; 2}} = {\sum\limits_{i = 1}^{Ncb}N_{t,i,{p\; 2}}}} & (3)\end{matrix}$The re-arranged bits can be used to fill up the time-frequencyresources, and the modulation positions in modulation symbols.

Step 2

Secondly, these bits are written into a three-dimensional space as shownin FIG. 14. These bits first fill up the space along the dimension offrequency (i.e. sub-carrier) index. Then they fill up the space alongthe dimension of time (i.e. OFDM symbol) index. At last they fill up thespace along the dimension of modulation position index. Note otherordering of dimensions is certainly possible and covered by the presentinvention. Each position in the three dimensional space may berepresented by a coordinate (b, t, f). If the first bit is placed at (0,0, 0), then the second bit should be placed at (0, 0, 1), the third bitshould be placed at (0, 0, 2), etc. After the frequency dimension isexhausted for a given OFDM symbol index, the OFDM symbol index isincreased. For example, the (M−1)-th bit should be placed at (0, 0,M−1), and the M-th bit may be placed at (0, 1, 0). After the frequencyand time indices are exhausted, the modulation position index isincreased. For example, the (MN−1)-th bit should be placed at (0, N−1,M−1), and the MN-th bit should be placed at (1, 0, 0). Note there mightbe some resource elements punctured or occupied by other channels andthus are not available to the data channel transmission. In a preferredembodiment of this invention, the time-frequency resources with aresource element map are represented in FIG. 15. The resource elementsassigned to a data transmission are grouped together to form a resourceelement map. The map shows the resource elements available to datatransmission and resource elements that are occupied by other channelssuch as reference signal, downlink control channels, etc. The resourceelements occupied by other channels are skipped. Note this resourceelement map can be reused for each modulation position, as shown in FIG.15. Eventually, the space that accommodates the coded bits can bedescribed as a cube as shown in FIG. 14, with some resource elementstaken by other channels and these taken resource elements are callednon-data resource elements.

Step 3

Thirdly, for each modulation position index and each OFDM symbol, thedata bits are interleaved along the frequency dimension. For example, abit-reverse-order (BRO) interleaver or a pruned bit-reverse-orderinterleaver may be used. Or any other interleaver may be used for thispurpose. Sometimes, one or multiple of simplified shuffling patterns maybe used. For example, cyclic shifts, or predeterminedinterleaving/re-arrangement/shuffling/swapping patterns may be used.These patterns may or may not change for each OFDM symbol and/or eachmodulation position index. Sometimes the number of resource elementsavailable in each OFDM symbol may be different due to different amountof puncturing or usage by other channels in these OFDM symbols. In thatcase, interleaver with different sizes may be used on different OFDMsymbols.

Step 4

Fourth, for each modulation position index and each sub-carrier, thedata bits are interleaved along the time dimension. For example, abit-reverse-order (BRO) interleaver or a pruned bit-reverse-orderinterleaver may be used. Or any other interleaver can be used for thispurpose. Sometimes, one or multiple of simplified shuffling patterns canbe used. For example, cyclic shifts, or predeterminedinterleaving/re-arrangement/shuffling/swapping patterns can be used.These patterns may or may not change for each modulation position and/orsub-carrier index. Sometimes, the number of resource elements availableon each sub-carrier index may be different due to different amount ofpuncturing or usage by other channels on this sub-carrier. In that case,interleaver with different sizes may be used on different sub-carriers.

Step 5

Fifth, for each sub-carrier and each OFDM symbol, the data bits areinterleaved along the dimension of modulation position index. Forexample, a bit-reverse-order (BRO) interleaver or a prunedbit-reverse-order interleaver may be used. Or any other interleaver canbe used for this purpose. Sometimes, one or multiple of simplifiedshuffling patterns can be used. For example, cyclic shifts, orpredetermined interleaving/re-arrangement/shuffling/swapping patternscan be used. These patterns may or may not change for each sub-carrierand/or each OFDM symbol. Preferred patterns will be explained later inthe present invention.

One preferred embodiment of the channel interleaver design consists atleast one of the above-stated five steps.

In the following description, embodiments and variations to theaforementioned steps are disclosed. Note these embodiments only describeone or multiple intermediate steps of the whole interleaving process.Particularly, pictorial illustrations used only show the effect of oneor multiple intermediate steps and may not reflect the final outcome ofthe interleaving process. For example, FIG. 19 shows the coded bits of acode block are spread over time, frequency, and spatial domain. Thecoded bits of the code block, however, are all in the first modulationposition of all the modulation symbols. Subsequent interleaving stepswill interleave these coded bits in time, frequency, and spatialdimension, and will shift the modulation positions of these coded bitsin modulation symbols.

Steps 1 and 2 attempt to spread the coded bits of each code block infrequency and time domains as much as possible to maximize frequency andtime diversity. For long term evolution (LTE) systems, frequencydiversity is normally more pronounced than time diversity within onetransmission. Therefore, it is preferable to first increase thefrequency sub-carrier index and then increase the OFDM symbol index.Different ordering of increasing the indices of different dimensions iscertainly covered by the present invention. Normally forward errorcorrection codes, especially when implemented with practical decoders,handle separated or random errors better than burst or contiguouserrors. Step 3 transfers burst errors in frequency domain into separatederrors. Step 4 transfers burst errors in time domain into separatederrors. For higher order modulation, each modulation position within amodulation symbol may enjoy different protection. Step 5 attempts torandomize or uniformly distribute the bits of each code block intodifferent modulation positions of modulation symbols so that on averagecoded bits of each code block enjoys the same level of protection. Theorder of Step 3, Step 4, and Step 5 may be changed without escaping theidea of the present invention. Certain steps may also be combined into asingle step. For example, Step 2 and Step 5 may be easily combined byjumping to a different modulation position as frequency and time indiceschange.

There are many alternative implementations to Steps 2, 3, and 4 thatachieve similar effects in spreading the coded bits of each code blocksinto time-frequency domain. In one embodiment of the invention, someother two-dimension matrices may be used to represent the time-frequencyresources instead of the resource element map. For example, atwo-dimension matrix with the number of rows equal to the number ofresource blocks and the number of columns equal to the number ofresource elements available for data in each resource block can be used.Suppose the data transmission is assigned as a number of N_(block)resource blocks, and there are a number of N_(DataRE) bits available fordata transmission in each resource block, then the coded bits may beplaced into the L×N_(block)×N_(DataRE) space. Preferably, the blockindex increases first, then the resource element index, then themodulation position index. By doing so, the adjacent coded bits areseparated into different resource blocks which are likely to experiencedifferent channel conditions. The same operation may also be describedas a row-column interleaving/permutation with a size ofN_(block)×N_(DataRE) applied on each modulation position, or as arow-column interleaving/permutation with the size ofN_(block)×N_(DataRE) applied on modulation symbols. On each modulationposition, the coded bits are written into a matrix N_(block)×N_(DataRE)with the block index increases first. Interleaving along the block indexor the data resource element (RE) index can certainly be performed, ifdesired. The purpose of interleaving along the block index is torandomize the location of coded bits into blocks that are far apart. Thepurpose of interleaving along the data RE index is to randomize thelocation of coded bits within a resource block. At last, when these bitsare mapped to time-frequency resources, they are read out and placed ontime-frequency resources with the resource element index increases firstto achieve the effect of row-column interleaving. Again, note thisoperation can be applied on the whole modulation symbols instead ofbeing applied on each modulation position.

Alternatively, to ease the implementation of Step 2, 3, and 4, onedimension may be employed to represent the time-frequency resources. Theresource elements are indexed on the resource element map. Theassignment of indices to resource elements may be arbitrary. Forexample, starting from the one with the lowest OFDM symbol index and thelowest sub-carrier index, the resource elements may be exhausted byfirst increasing the sub-carrier index and then increasing the OFDMsymbol index. Alternatively, the resource elements may be exhausted byfirst increasing the OFDM symbol index and then increasing thesub-carrier index. One example is shown in FIG. 17 (a) and FIG. 17 (b).FIG. 17 (b) shows a time (OFDM symbol index)-frequency (sub-carrierindex) space for accommodating data coded bits, this time-frequencyspace is shown as one dimension in FIG. 17 (a), and this dimension growsalong modulation position index direction and becomes a two-dimensionalmatrix as shown in FIG. 17 (a). The resources elements occupied by dataas shown in FIG. 17 (b), i.e., resources elements 1-32 are expended inthe modulation position index and becomes ha two-dimensional matrix asshown in FIG. 17 (a). After the time-frequency space as shown in FIG. 17(b) is filled up by the coded bits of multiple resource blocks i, j,interleave may be done along the resource element dimension to spreadadjacent coded bits in the time-frequency resources. Bit-reverse-order,or pruned bit-reversal-order, or any other type of interleaver may beused.

Alternatively, the implementation of Step 1, 2, 3, and 4 may be moreintegrated with the encoding processes ahead of these steps. Forexample, the HSDPA system assumes the systematic bits of all code blocksare placed together followed by the parity 1 bits of all coded blocksand then followed by the parity 2 bits of all coded blocks. To group thesystematic bits, parity 1 bits, and parity 2 bits of at least one codeblock after rate matching, this can also be achieved by performing theentire rate matching process separately for the code blocks. FIG. 18shows an implementation of rate matching and bit collection on a codeblock basis. Each code block, i.e., code block 1, code block 2, . . . ,code block Ncb, is processed by bit separator 610, first rate matchingstage 611, virtual IR buffer 613, second rate matching stage 615 and bitcollection stage 617 in a sequence. All of the re-arranged bits thenenter a stage 915, which is called channel interleaver stage. For eachcode block, the processes are similar to the processes taught in thedescriptions of FIG. 10. Therefore, detailed explanation is omittedhere. In this implementation, the encoder output of at least one codeblock are passed through first rate matching stage 611, virtual IRbuffer 613, and second rate matching stage 615 separately. So, theoutput of the second rate matching will naturally have the systematicbits, parity 1 bits, and parity 2 bits of the code block groupedtogether. Although the actual rate matching processes for multiple codeblocks is done separately, the parameter and configuration of these ratematching processes for multiple code blocks may need coordination.Certainly, this invention covers other variants of the implementationwith some steps in the rate matching process simplified, modified, orskipped, as long as the rate matching process is done on code blockbasis. For example, the first rate matching stage 611, virtual IR buffer613, and second rate matching process 615 may be simplified and combinedinto one step which simply selects the appropriate coded bits for eachtransmission. For example, bit separation stage 610 and bit collectionstage 617 may not be necessary if the encoder output is already groupedin to systematic bits, parity 1 bits, and parity 2 bits.

For a suboptimal though simpler implementation, Step 1 may be skipped.In this case, the systematic bits, parity 1 bits, and parity 2 bits ofeach code block are not grouped together. With the efforts in the restof the interleaving steps, the coded bits of each code block are stillsufficiently spread and good performance may be achieved.

The aforementioned embodiments of channel interleaver design may beeasily extended to the case of MIMO transmissions. Suppose multiplelayers are allocated to a MIMO codeword. This scenario may apply to longterm evolution (LTE) systems, e.g., when the SU-MIMO transmission hasrank greater than 1. In this case, a spatial dimension is added in thechannel interleaver design. The space for the coded bits may bedescribed as a four-dimensional space in time, frequency, space andmodulation positions. To illustrate the idea in a three-dimensionalspace, which allows us a pictorial presentation, the time-frequencydimension is simplified into one dimension of resource elements as shownin FIG. 17. Therefore, the space for coded bits may be represented as athree-dimensional space in resource elements, space, and modulationpositions, as shown in FIG. 19. FIG. 19 illustrates spreading coded bitsof each code clock over time, frequency and spatial domain. Theinterleaver will allocate coded bits of each code block along the spacedimension first, then along the resource element dimension to make surethe code block collects maximal diversity in time, frequency, and space.Resource element index refers resource elements dimension, spatialdimension refers to space dimension, and modulation position indexrefers to modulation positions dimension. If multiple code blocks aretransmitted, the code blocks within each of the codeword should bespread over time, frequency, and spatial domain. In FIG. 19, coded bits1, 2, 3, . . . , 16 belong to one code block while coded bits 1′, 2′,3′, . . . , 16′ belong to another code block.

In another embodiment of the invention, the aforementioned embodimentsare extended to MIMO transmissions with different spatial dimensions ondifferent resource elements.

In a MIMO system, the rank (number of spatial dimensions, or layers) maybe different on different frequency resource elements. Theaforementioned embodiments may also be extended to transmissions withdifferent modulation order on different resources. For example, if tworesource blocks have very different CQI, the transmitter may usedifferent modulation orders on these two resource blocks. In this case,the design goal of spreading coded bits of each code block as much aspossible over time, frequency, spatial, and modulation positions stillapplies. Special handling needs to be implemented to handle the case ofdifferent spatial dimensions or different modulation orders on differenttime-frequency resources. For example, similar to the resource elementmap, a map can be constructed to include spatial and modulation positiondimension. The layers or the modulation positions that are not availablewill be skipped. FIG. 20 shows an example of writing coded bits intoresources with different layers and different modulation orders. In FIG.20, different modulation orders such as QPSK, 16-QAM and 64-QAM areshown, and rank 2 on resource element index 0,1,2,3,10,11,12,13,14, and15 and rank 1 on resource element index 4,5,6,7,8 and 9. In thisexample, the two code blocks still try to spread over spatial dimensionsfirst, when the spatial dimension is collapsed to 1, as in resourceelements 4, 5, 6, 7, 8, 9, the two code blocks will both be placed inthe same layer at the spatial dimension. Each code block, however, willstill be spread in time and frequency dimension (Not shown in FIG. 20because time and frequency dimensions are shown as one dimension ofresource elements). After all coded bits are mapped into resourceelements, other interleaving processing such as row-column interleaving,interleaving of modulation positions, can be performed to furtherrandomize the location of coded bits.

A pictorial illustration of channel interleaver when differentmodulation orders are used on different resources is also shown in FIG.21. FIG. 21 illustrates channel interleaver with different modulationorder on resources. In this case, resource block A uses 16-QAM whileresource block B uses 64-QAM. The coded bits fill up the space definedby time, frequency, and available modulation positions on each resourceelements, skipping resource elements occupied by other channels. Insummary, the aforementioned interleaving steps and embodiments apply inthis case.

In another embodiment of the invention, systematic bits priority isgiven in mapping coded bits, and modulation symbols formed by thesecoded bits, onto resource elements and spatial dimensions. FIG. 22 showsan example of spread coded bits on resources with different spatialdimensions. For example, as shown in FIG. 22, the rank (number ofspatial dimensions, or layers) is 2 on resource element 0, 1, 2, 3, 10,11, 12, 13, 14, 15; and the rank is 1 on resource element 4, 5, 6, 7, 8,9. In this example, the same modulation order is applied to all theresources and all the layers, and the modulation order is 16-QAM. Due tothe interference between MIMO layers, the CQI (channel qualityindicator) on resource element 4, 5, 6, 7, 8, 9 is often higher than theCQI per layer on resource elements 0, 1, 2, 3, 10, 11, 12, 13, 14, 15.In this case, the systematic bits are given more protection by beinggiven priority to be placed on those resource elements with smallernumber of layers. On the other hand, parity bits are given priority tobe placed on those resource elements with bigger number of layers. Inthe example shown in FIG. 22, all systematic bits, i.e., S₀, S₁, S₂, S₃,S₄, S₆, S₇, S₈, S₉, are placed on resource elements index 4, 5, 6, 7, 8,9, while all parity bits, i.e., P_(0,0), P_(1,1), P_(0,2), P_(1,3),P_(1,0), P_(0,1), P_(1,2), P_(0,3), P_(0,4), P_(1,5), P_(0,6), P_(1,7),P_(0,8), P_(1,9), P_(1,4), P_(0,5), P_(1,6), P_(0,7), P_(1,8), P_(0,9),are placed on resource elements 0, 1, 2, 3, 10, 11, 12, 13, 14, 15.

The prioritization of systematic bit may also be implemented by definingmultiple regions along the dimension of modulation positions. Forexample, for the 64-QAM constellation as defined in Table 2 and FIG. 12,two regions are defined, i.e., a first region which contains b0, b1, b2,b3 for systematic bits and a second region which contains b4, b5 forparity bits. Systematic bits are prioritized in the first region whileparity bits are prioritized in the second region. The first region maycontain some parity bits in certain cases, e.g., there are not enoughsystematic bits to fill up the first region. Similarly, the secondregion may contain some systematic bits in certain cases, e.g., thereare not enough parity bits to fill up the second region. All theaforementioned embodiments, interleaving steps may be performedseparately in these two regions. Because the modulation positions aredivided into two regions, the interleaving/permutation along modulationpositions needs to be done separately for these two regions. In otherwords, the region {b0, b1, b2, b3} is permuted as if it is a 16-QAMmodulation while {b4, b5} is permuted as if it is a QPSK modulation.Again, there may still be variations of this idea. For example, insteadof defining two regions, two starting points and directions may bedefined separately for systematic bits and parity bits. Systematic bitsstart at the modulation positions with the strongest protection and movetowards modulation positions with weaker protection while parity bitsstart at the modulation positions with the weakest protection and movetowards modulation positions with stronger protection.

In another embodiment of the present invention, the coded bits of eachcode block are allocated as uniformly as possible on differentmodulation positions. There are various ways to achieve this goal. Oneapproach is to enumerate all of the permutation patterns of themodulation positions. The permutation patterns for the modulationpositions of QPSK and 16-QAM are shown in Table 3. By assigningdifferent permutation patterns to different modulation symbols, themodulation positions of the coded bits are changed in a code block. Bydoing so, the coded bits of each code block are roughly uniformlydistributed among all modulation positions. So no code block isparticularly favored or disadvantaged. Another benefit of interleavingor permutation along the modulation positions is to alleviate theproblem caused by I-Q imbalance. If the SNR on I-branch and Q-branch isdifferent, interleaving or permutation can ensure the coded bits of eachcode block spread across both I and Q branches.

TABLE 3 Modulation Modulation position permutation QPSK 01, 10 16-QAM0123, 0132, 0213, 0231, 0312, 0321, 1023, 1032, 1230, 1203, 1302, 1320,2013, 2031, 2103, 2130, 2301, 2310, 3012, 3021, 3102, 3120, 3201, 3210

Obviously, a subset of the permutation patterns may be selected. Forexample, one seed permutation pattern with its cyclic shifted versionsmay be used as one subset of patterns. A few examples for QPSK, 16-QAM,and 64-QAM are shown in Table 4. In Table 4, the natural orderingpatterns are used as the seed pattern. The subset of permutationpatterns are generated by cyclic shifts of the seed pattern. Thesepermutation patterns may be applied to modulation symbols on differentresource elements. These subsets of patterns may change every resourceelements, or every a few resource elements. By doing so, the coded bitsof each code block will be shifted to different modulation positions indifferent modulation symbols. Therefore, each code block will getroughly equal protection from modulation. This technique may also beapplied in retransmissions in HARQ. One example of application is tochange the permutation patterns of the same modulation symbols acrosstransmissions. This may be achieved by using different cyclic shifts ofthe same seed permutation pattern, or using different seed permutationpatterns in retransmissions.

TABLE 4 Seed permutation Modulation pattern Cyclic shifts of seedpermutation patterns QPSK 01 01, 10 16-QAM 0123 0123, 1230, 2301, 301264-QAM 012345 012345, 123450, 234501, 345012, 450123 501234

Certainly, the selection of a subset of permutation patterns may bevarious and depends on other design objectives. For example, not allcyclic shifts are needed in the selected subset. Cyclic shifts frommultiple seed permutation patterns may be selected. In one preferableembodiment of the invention, the preferred seed for QPSK, 16-QAM (asshown in FIG. 11), 64-QAM (as shown in FIG. 12) are listed in Table 5.Because of the relatively higher order modulation, the protection levelon some positions is equal, while the protection level on some otherpositions is different. For the constellation of 16-QAM as shown in FIG.11 and the constellation of 64-QAM as shown in FIG. 12, b0 and b1receives the strongest protection, b2 and b3 receive less protection,while b4 and b5 (in the case of 64-QAM) receive the least amount ofprotection. According to Table 5, 0213 and its cyclic shifted versionsare used for 16-QAM, while 042153 and its cyclic shifted versions areused for 64-QAM. Another preferred seed permutation for 16-QAM is 0312(not shown in Table 5). Other preferred seed permutations for 64-QAM are024135, 052143, 043152, 053142, 025134, 034125, 035124 (not shown inTable 5). The preferred seed permutation pattern may change as theconstellation design of 16-QAM or 64-QAM changes.

TABLE 5 Preferred Seed permutation Modulation pattern Cyclic shifts ofseed permutation patterns QPSK 01 01, 10 16-QAM 0213 0213, 2130, 1302,3021 64-QAM 042153 042153, 421530, 215304, 153042, 530421, 304215

FIG. 23 shows a general method of obtaining the preferred permutationpattern for 64-QAM. The modulation positions are placed with the samelevel of protection at two ends of a diameter of a circle, and themodulation positions are placed with different level of protection atdifferent angle along the circle. For example, the modulation positionsof b0 and b1 are placed with the strongest protection at two ends ofdiameter A-A′ of the circle as shown in FIG. 23, the modulationpositions of b2 and b3 are placed with the weaker protection at two endsof diameter B-B′ of the circle, and the modulation positions of b4 andb5 are placed with the weakest protection at two ends of diameter C-C′of the circle. Different preferred seed permutation patterns, and theircyclic shifts, may be obtained by reading the positions along thecircle, starting from any position and by going either clockwise orcounter-clockwise. In this way, maximum separation of the modulationpositions with the same level of protection is achieved. This method isalso applicable to other modulation orders. As shown in Table 5, apreferred seed for 64-QAM is 042153. According to FIG. 23, when startingfrom point A′ and counting in counter-clockwise along the periphery ofthe circle, b0b4b2b1b5b3 is achieved. Therefore, a preferred seed for64-QAM is 046153. By the same method, all of the preferred permutationpattern maybe achieved for 64-QAM. Although the seed permutationpatterns are generated in this way, and their cyclic shifts, arepreferable, this invention certainly covers the application of themodulation position interleaving, permutation, shuffling, swapping,re-arranging on resource elements and/or across retransmissions with anypattern or in any fashion.

In another embodiment of this invention, an iterative operation isproposed for receiving the multiple code blocks that are multiplexedtogether within modulation symbols. With the aforementioned channelinterleaver design, the coded bits of different code blocks aremultiplexed in the same modulation symbol. FIG. 24 illustrates aniterative receiver for decoding multiple code blocks multiplexed withinsame modulation symbols. Here, an iterative operation is proposed toimprove the receiver performance. An illustration of this operation isshown in FIG. 24. After processed by receiver front end and somebaseband processing stage 690, e.g., FFT, channel estimation,equalization etc., soft values of coded bits are obtained bydemodulating the modulation symbol by demodulator 692. These soft valuesare then deinterleaved by deinterleaver 694 and are fed into decoder696. There are multiple code blocks. Decoder 696 attempts to decode one,or multiple, or all of the code blocks. Parallel processing is alsopossible in the decoding operation. After the decoding operation, somecode blocks may be successfully decoded while some others are not. Inthis case, the code blocks of those decoded code blocks arereconstructed. Because the coded bits of these blocks are multiplexed inthe same modulation symbols with the coded bits of those code blocksthat are unsuccessful, the information of these coded bits are used tohelp the detection of the coded blocks that are yet unsuccessful. Codeclocks successfully decoded are feedback to an encoder 698 and then fedto an interleaver 699. Therefore, the information of these successfullydecoded code blocks are used to help the detection of the coded blocksthat are yet unsuccessful.

FIG. 25 shows an example of a reduced constellation which improves thedetection performance of the transmission. For example, a 16-QAMconstellation b3b2b1b0 is shown in sub-figure (a) of FIG. 24. b3 issupposed to belong to code block 1, b2 and b1 are supposed belong tocode block 2, and b0 is supposed belong to code block 3. If code block 2is successfully decoded, the knowledge of the value of b2 and b1 isobtained. If b2=0 and b1=1, then the constellation is reduced as shownin FIG. 25. In this case, the demodulation of b3 and b0 based on thereduced constellation may have an improved performance.

In another embodiment of the invention, the iterative operation may beperformed without correctly decoding and re-encoding some of the codeblocks. Instead, reliability of the coded and information bits may beused to pass through the iterations to improve detection performance.One representation of reliability is called extrinsic information, whichis the new likelihood information about each bit that is passed betweenthe multiple processing blocks within the iterative loop. An example isshown in FIG. 26. FIG. 26 illustrates an alternative iterative receiverfor decoding multiple code clocks multiplexed in same modulationsymbols. Because FIG. 26 is almost same as FIG. 24, detailed explanationis omitted and only difference will be described. Extrinsic informationis passed between demodulator 692 and decoder 696. Each takes theextrinsic information from the other as prior information in thecalculation of likelihood of each bit and generates a new round ofextrinsic information. For a successful decoding, as the iteration goeson, the likelihood of the bits will improve and eventually leads todecoding success.

In a long term evolution (LTE) system, the downlink subframe structureis shown in FIG. 27.

As shown in FIG. 27, each subframe contains two slots with each slotcontaining seven OFDM symbols (i.e. OFDM symbols 0-6) in time domain.Control channel signals are located in the first two or three OFDMsymbols in a subframe. In this case, control channel signals are locatedin the first two OFDM symbols. Reference signals are located at OFDMsymbol 0, 4, 7, and 11. For simplicity, only the reference signals ofthe first transmit antenna will be discussed. In frequency domain, datamay be presented by multiple resource blocks, such as resource blocks iand j. The ideas of the present invention may certainly be extended tosystems with multiple transmit antennas and multiple reference signals.In order to maintain good channel estimation performance, interpolationor averaging of downlink reference signals is usually used. For example,as shown in FIG. 27, interpolation of reference signals at resourceelements A, B, C, and D can be used to obtain channel estimation forresource element S with improved performance. However, this also meansthat the demodulation of the modulation symbol in resource element Sneed to wait until reference signals in resource element C and D arereceived. In other words, if the demodulation of resource element Shappens before OFDM symbol 11 that contains resource elements C and D,the channel estimation performance for resource element S may benegatively impacted.

In the present invention, methods and apparatus to enable fast decodingof transmissions with information bits or parity bits from multiplecoded packets are also proposed.

Aspects, features, and advantages of the present invention are readilyapparent from the following detailed description, simply by illustratinga number of particular embodiments and implementations, including thebest mode contemplated for carrying out the invention. The presentinvention is also capable of other and different embodiments, andseveral details can be modified in various obvious respects, all withoutdeparting from the spirit and scope of the invention. Accordingly, thedrawings and description are to be regarded as illustrative in nature,and not as restrictive. The present invention is illustrated by way ofexample, and not by way of limitation, in the figures of theaccompanying drawings. In the following illustrations, downlink datachannel in long term evolution (LTE) systems is used as an example. Thetechnique illustrated in the present invention, may certainly be used inuplink data channel in long term evolution (LTE) systems, controlchannels in either downlink or uplink in long term evolution (LTE)systems, and other data, control, or other channels in other systemswhenever applicable.

When multiple code blocks are transmitted, the performance of the datatransmission is dictated by the code block which has the worstperformance. The channel interleaver, including the mapping from codedbits of different code blocks to modulation symbols, and the mappingfrom modulation symbols to time, frequency, and spatial resources, needsto be carefully designed to make sure each code block get roughly thesame level of protection. When multiple code blocks are transmitted, itis beneficial to allow the receiver to start decoding of some codeblocks while the receiver is still demodulating modulation symbols forother code blocks. In a long term evolution (LTE) system, this presentsa challenge because the channel estimation performance might be impactedif there are not enough reference signals at the time of demodulationand decoding. In the present invention, techniques that allow fastdecoding of multiple code blocks are proposed while good channelestimation performance is maintained.

In an OFDMA system, e.g., long term evolution (LTE), there are normallymultiple OFDM symbols in a subframe. In long term evolution (LTE),reference signals are present in a few OFDM symbols called ReferenceSignal OFDM symbols, while absent in other OFDM symbols.

In one embodiment of the present invention, multiple OFDM symbols in asubframe are separated into groups with the boundary between at leasttwo groups located in the Reference Signal OFDM symbols, or those OFDMsymbols right before or right after the Reference Signal OFDM symbols.Each group contains resource elements that will carry coded bits from atleast one code block. The resource elements in each group are contiguousor closely to each other in time domain. The receiver can start decodingof at least one code block after receiving all the resource elements ineach group.

FIG. 28 shows one example of the configuration of grouping multiple OFDMsymbols in a subframe constructed according to one embodiment of thepresent invention.

As shown in FIG. 28, control channel signals are located in the firsttwo or three OFDM symbols in a subframe. In this case, control channelsignals are located in the first two OFDM symbols. Group 1 is defined asa set of resource elements in OFDM symbol 2, 3, and 4 that are availableto data channel; Group 2 is defined as a set of resource elements inOFDM symbol 5, 6 and 7 that are available to data channel; Group 3 isdefined as a set of resource elements in OFDM symbol 8, 9, and 10 thatare available to data channel; Group 4 is defined as a set of resourceelements in OFDM symbol 11, 12, and 13 that are available to datachannel. OFDM symbol 4 contains downlink reference signal; the boundarybetween Group 1 and Group are located between OFDM symbol 4 and 5. OFDMsymbol 11 contains downlink reference signals; the boundary betweenGroup 3 and Group 4 are located between OFDM symbol 10 and Because theboundary is always at or close to where the reference signal is, thereceiver can start decoding of the code blocks carried within a groupright after the group is received, or wait for one additional OFDMsymbol, without sacrificing channel estimation performance. The receivercan take advantage of the latest available reference signal for thedemodulation and decoding of the code blocks carried within a group. Forexample, there are 8 code blocks, the coded bits of two code blocks maybe put into each group. For example, coded bits of code block 1 and 2are contained in Group 1. After receiver received OFDM symbol 2, 3, 4,because all the resource elements in Group 1 is contained in these OFDMsymbols, the receiver has received all the coded bits for code block 1and 2. Therefore, the receiver can start decoding of these two codeblocks. In this way, the receiver does not have to wait until the end ofthe subframe (after OFDM symbol 13) to start decoding. This designbrings a few benefits to the receiver design in terms of hardwarecomplexity and power consumption.

FIG. 28( a) is a flow chart illustrating a method of transmitting datasignals by separating resource elements having coded bits suitable forthe practice of the principles of one embodiment of the presentinvention. FIG. 28 (b) is a flow chart illustrating a method ofreceiving and decoding grouped resource elements having coded bits at areceiver. In FIG. 28( a), data bits are modulated in step 911, are thentransformed from serial to parallel in step 912. Then data istransformed by IFFT method at step 915 and then processed from parallelto serial at step 916. Then the data in different code blocks are mappedto resources elements that are separated into different groups in step917. Finally, transmitting front end having one or multiple transmittingantennas transmits the OFDM symbols that includes one or multiple groupsof resource elements. In FIG. 28( b), the receiver starts receive OFDMsymbols that include one group of resource elements by receiving frontend process at step 951. Then the received OFDM symbols are processed byserial to parallel stage at step 952 and by FFT method at step 953. Thegroup of resource elements are processed by parallel to serialprocessing at step 956 and finally are demodulated at step 957. Thereceiver then decodes the coded bits of the resources elements withinthe one group at step 958. As the receiver continues to receive OFDMsymbols, the following groups of resource elements can be received andprocessed.

FIG. 29 shows another example of the configuration of grouping multipleOFDM symbols in a subframe suitable for the practice of the principlesof the present invention. In this example, two groups are defined.Control channel signals are located in the first two or three OFDMsymbols in a subframe. Group 1 includes resource elements in OFDM symbol2, 3, 4, 5, 6, 7 and Group 2 includes resource elements in OFDM symbol8, 9, 10, 11, 12, 13. OFDM symbols 2-13 in a subframe are separated intotwo groups with the boundary between two groups located at OFDM symbol 7and 8. Note that OFDM symbol 7 carries Reference Signals. Differentconfiguration of groups can be used in different situations, such as,but not limited to, different UEs, different subframes, differentquality of service, etc. without departing from the spirit of thisinvention.

FIG. 30 shows another example of the configuration of grouping multipleOFDM symbols in a subframe suitable for the practice of the principlesof the present invention. In this example, although each group includesresource elements from contiguous OFDM symbols, some OFDM symbols, e.g.,OFDM symbols 5, 8, and 11, may contain multiple groups. Again, note thatthe boundaries between groups are all located in OFDM symbols that carryReference Signals or OFDM symbols that are immediately before or afterthe OFDM symbols that carry Reference Signals. This design allows moreflexible group definition than the OFDM symbol based grouping whilemaintaining the benefits of allowing fast decoding without channelestimation performance loss.

In another embodiment of this invention, groups are defined based oncode blocks instead of resource elements. FIG. 32 shows an example ofthe configuration of grouping code bock suitable for the practice of theprinciples of this embodiment of the present invention. Each groupcontains coded bits of at least one code block and may contain multiplecode blocks. For example, code blocks 1 and 2 may be grouped as thefirst group, code blocks 3 and 4 may be grouped as the second group,code blocks 5 and 6 may be grouped as the third group, and code blocks 7and 8 may be grouped as the fourth group. The first group is placed inthe first a few OFDM symbols, and the second group is placed in the nexta few OFDM symbols, etc. In this way, fast decoding of some code blockswithout waiting until the end of a subframe is allowed.

With the group defined in aforementioned embodiments, either based onresource elements or code blocks, the rest of channel interleavingoperations within each group may be defined. The channel interleaver maybe quite general. For example, the channel interleaver may spread thecoded bits of each code block within a group to as many resourceelements in this group as possible. The channel interleaver may spreadthe coded bits of each code block within a group to different modulationpositions as evenly as possible. The channel interleaver may attempt tomake sure each modulation symbol within a group contains coded bits frommultiple code blocks so that burst error on modulation symbols is spreadon these code blocks.

The aforementioned embodiments of channel interleaver design can beextended to the case of MIMO transmissions. Suppose multiple layers areallocated to a MIMO codeword. This scenario can happen in long termevolution (LTE) systems, e.g., when the SU-MIMO transmission has rankgreater than 1. In this case, a spatial dimension can be added to thedefinition of a group. A Multi-Input-Multi-Output (MIMO) codewordrepresents a transport block. A Multiple Input Multiple Output (MIMO)processor generating soft bits for at least one of a plurality of codeblocks of a Multiple Input Multiple Output (MIMO) codeword

FIG. 31 shows examples of parallel processing for successiveinterference cancellation either with or without group cyclic delaydiversity (CRC) suitable for the practice of the principles of thepresent invention.

A shown in FIG. 31, four groups are defined within a subframe: Group 1can be defined as a set of resource elements in OFDM symbol 2, 3, and 4,including multiple MIMO layers or MIMO streams on those resourceelements; Group 2 can be defined as a set of resource elements in OFDMsymbol 5, 6, and 7, including multiple MIMO layers or MIMO streams onthose resource elements; etc.

As shown in subfigure (a) of FIG. 31, within each group there are twolayers; within each group there are two layers or streams; within eachMIMO layer or MIMO stream there are four groups. In a multi-codewordMIMO transmission, each layer may incorporate a corresponding MIMOcodeword (CW), i.e. CW1 and CW2, and each CW carries multiple codeblocks and a 24-bit CRC as shown in subfigure (b) of FIG. 31. For eachMIMO codeword, this CRC is applied to the whole MIMO codeword (atransport block having multiple code blocks), i.e., all the code blocksin Group 1, 2, 3, and 4 that belong to the MIMO codeword. Therefore,with the group definition, decoding of code blocks in MIMO CW1 may bestarted immediately after demodulation of modulation symbols in Group 1.By doing so, the demodulation of the later groups is parallelized withthe decoding of earlier groups. In addition, with the help of this CRC,the interference from CW1 to CW2 may be cancelled by successiveinterference cancellation.

Further, the parallel processing capability may be significantlyenhanced. In another embodiment of the present invention, CRC is addedto one or multiple code blocks of a codeword within a group. One exampleis shown in subfigure (c) of FIG. 31: one CRC may be attached to each ofgroups. With the group definition, decoding of code blocks in CW1 may bestarted immediately after demodulation of Group 1. By doing so, thedemodulation of the later groups is parallelized with the decoding ofprevious groups. With the per-group CRC, the interference from CW1 toCW2 in Group 1 may be cancelled and decoding of code blocks of CW2 inGroup 1 may be started immediately after the decoding of those codeblocks of CW1 in Group 1. By doing so, the demodulation of the latergroups in CW1, the decoding of earlier groups in CW1, the successiveinterference cancellation, the demodulation of the later groups in CW2,and the decoding of earlier groups in CW2 can all be processedparalleled in one way or another.

Certainly, CRC may be added to the groups of both MIMO codewords CW1 andCW2 separately. In that case, it enables parallel processing even in thecase of an iterative receiver. In other words, when the iterativereceiver is used to decode CW1 and CW2, the iterative receiver for Group1 is used in the decoding of the code blocks of CW1 in Group 1, and thedecoding of the code blocks of CW2 in Group 1. In this case, even theiterative receiver can be parallelized between groups.

Certainly, many variations and receiver structures may be obtained. Forexample, the group definition across these two layers needs not to beexactly synchronized. This may cause some delay in processing or someperformance degradation, but may allow more flexibility in groupdefinition. Even the number of groups within each layer may bedifferent.

1. A transmitter, comprising: a first cyclic redundancy check (CRC)calculator configured to generate a transport block CRC for at least onetransport block of information bits; a separator configured to segmentthe transport block into a plurality of code blocks; a second CRCcalculator configured to generate a plurality of code block CRCs for theplurality of code blocks, wherein at least one of the code block CRCs isgenerated based upon a corresponding code block; and a processing unitconfigured to transmit the plurality of code blocks and the plurality ofcode block CRCs.
 2. The transmitter of claim 1, further comprising: anencoder configured to encode bits in each code block within theplurality of code blocks using a forward error correcting code, whereinthe second CRC generator is configured to generate the code block CRCsbased upon the encoded code blocks.
 3. The transmitter of claim 1,wherein each of the plurality of code block CRCs is generated based upona corresponding one of the plurality of code blocks.
 4. The transmitterof claim 1, wherein each of the plurality of code block CRCs isgenerated based upon at least the corresponding one of the plurality ofcode blocks.
 5. The transmitter of claim 1, wherein at least one of theplurality of code block CRCs is generated based upon a combination ofthe corresponding code block and at least one code block preceding thecorresponding code block.
 6. The transmitter of claim 1, wherein theplurality of code blocks comprise at least one code block from which nocode block CRC is generated.
 7. The transmitter of claim 1, furthercomprising: an encoder configured to encode data bits of the pluralityof code blocks using a type of forward error correcting code.
 8. Thetransmitter of claim 1, wherein the first CRC generator is furtherconfigured to attach the transport block CRC to the transport block, andwherein the second CRC generator is further configured to attach the atleast one code block CRC to the corresponding code block.
 9. Thetransmitter of claim 1, wherein the first CRC generator is configured togenerate the transport block CRC for the transport block before thetransport block is segmented into the plurality of code blocks.
 10. Thetransmitter of claim 1, wherein each of the plurality of code block CRCsare generated based on a different one of the plurality of code blocks.11. The transmitter of claim 1, wherein each of the code block CRCs isgenerated based upon a corresponding one of the plurality of codeblocks.
 12. The transmitter of claim 1, wherein each of the code blockCRCs is generated based upon at least the corresponding one of theplurality of code blocks.
 13. The transmitter of claim 12, wherein theeach of the code block CRCs is generated based upon a combination of thecorresponding of code block and code blocks preceding the correspondingcode block.
 14. A method for transmitting data bits, the methodcomprising: generating a transport block cyclic redundancy check (CRC)for at least one transport block of information bits; segmenting thetransport block into a plurality of code blocks; generating a pluralityof code block CRCs for the plurality of code blocks, wherein at leastone of the code block CRCs is generated based upon a corresponding codeblock; and transmitting the plurality of code blocks and the pluralityof code block CRCs.
 15. The method of claim 14, further comprising:encoding bits in a code block selected from the plurality of code blockusing a forward error correcting code; and generating the code blockCRCs based upon the encoded code blocks.
 16. The method of claim 14,wherein each of the plurality of code block CRCs is generated based upona corresponding one of the plurality of code blocks.
 17. The method ofclaim 14, wherein each of the plurality of code block CRCs is generatedbased upon at least the corresponding one of the plurality of codeblocks.
 18. The method of claim 14, wherein at least one of theplurality of code block CRCs is generated based upon a combination ofthe corresponding code block and at least one code block preceding thecorresponding code block.
 19. The method of claim 14, wherein theplurality of code blocks comprise at least one code block from which nocode block CRC is generated.
 20. The method of claim 14 furthercomprising: encoding data bits of the plurality of code blocks using atype of forward error correcting code.
 21. The method of claim 14,further comprising: attaching the transport block CRC to the transportblock; and attaching the at least one code block CRC to thecorresponding code block.
 22. The method of claim 14, wherein thetransport block CRC for the transport block is generated before thetransport block is segmented into the plurality of code blocks.
 23. Themethod of claim 14, wherein each of the plurality of code block CRCs aregenerated based on a different one of the plurality of code blocks. 24.The method of claim 14, wherein each of the code block CRCs is generatedbased upon a corresponding one of the plurality of code blocks.
 25. Themethod of claim 14, wherein each of the code block CRCs is generatedbased upon at least the corresponding one of the plurality of codeblocks.
 26. The method of claim 25, wherein the each of the code blockCRCs is generated based upon a combination of the corresponding of codeblock and code blocks preceding the corresponding code block.